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LT3573 Isolated Flyback Converter Without an Opto-Coupler FEATURES
n n n n n n n n n n
DESCRIPTION
The LT(R)3573 is a monolithic switching regulator specifically designed for the isolated flyback topology. No third winding or optoisolator is required for regulation. The part senses the isolated output voltage directly from the primary side flyback waveform. A 1.25A, 60V NPN power switch is integrated along with all control logic into a 16lead MSOP package. The LT3573 operates with input supply voltages from 3V to 40V, and can deliver output power up to 7W with no external power devices.The LT3573 utilizes boundary mode operation to provide a small magnetic solution with improved load regulation.
LT, LTC and LTM are registered trademarks of Linear Technology Corporation. All other trademarks are the property of their respective owners. Patents pending.
3V to 40V Input Voltage Range 1.25A, 60V Integrated NPN Power Switch Boundary Mode Operation No Transformer Third Winding or Optoisolator Required for Regulation Improved Primary-Side Winding Feedback Load Regulation VOUT Set with Two External Resistors BIAS Pin for Internal Bias Supply and Power NPN Driver Programmable Soft-Start Programmable Power Switch Current Limit Thermally Enhanced 16-Lead MSOP
APPLICATIONS
n n
Industrial, Automotive, Medical Low Power PoE
TYPICAL APPLICATION
5V Isolated Flyback Converter
VIN 12V TO 24V B340A VIN SHDN/UVLO 51.1k RFB RREF TC RILIM SS VC 28.7k 10k 10nF 20k 1nF
3573 TA01
Load Regulation
3
24H 1N4148 80.6k
2.6H
47F VOUT
-
OUTPUT VOLTAGE ERROR (%)
10F
357k
3:1 0.22F 2k
VOUT+ 5V, 0.7A
2 1 0 -1 -2 -3 0 200 400 600 800 1000 1200 1400
3573 TA01b
VIN = 24V VIN = 12V
LT3573
6.04k SW GND TEST BIAS
IOUT (mA)
4.7F
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LT3573 ABSOLUTE MAXIMUM RATINGS
SW ............................................................................60V VIN, SHDN/UVLO, RFB, BIAS .....................................40V SS, VC, TC, RREF , RILIM ...............................................5V Maximum Junction Temperature........................... 125C Operating Junction Temperature Range (Note 2) LT3573E .............................................-40C to 125C Storage Temperature Range...................-65C to 150C
PIN CONFIGURATION
TOP VIEW GND TEST GND SW VIN BIAS SHDN/UVLO GND 1 2 3 4 5 6 7 8 16 15 14 13 12 11 10 9 GND TC RREF RFB VC RILIM SS GND
17
MSE PACKAGE 16-LEAD PLASTIC MSOP TJMAX = 125C, JA = 50C/W, JC = 10C/W EXPOSED PAD (PIN 17) IS GND, MUST BE CONNECTED TO GND
ORDER INFORMATION
LEAD FREE FINISH LT3573EMSE#PBF LT3573IMSE#PBF TAPE AND REEL LT3573EMSE#TRPBF LT3573IMSE#TRPBF PART MARKING* 3573 3573 PACKAGE DESCRIPTION 16-Lead Plastic MSOP 16-Lead Plastic MSOP TEMPERATURE RANGE -40C to 125C -40C to 125C
Consult LTC Marketing for parts specified with wider operating temperature ranges. *The temperature grade is identified by a label on the shipping container. For more information on lead free part marking, go to: http://www.linear.com/leadfree/ For more information on tape and reel specifications, go to: http://www.linear.com/tapeandreel/
ELECTRICAL CHARACTERISTICS
PARAMETER Input Voltage Range Quiescent Current Soft-Start Current SHDN/UVLO Pin Threshold SHDN/UVLO Pin Hysteresis Current Soft-Start Threshold Maximum Switching Frequency Switch Current Limit Minimum Current Limit Switch VCESAT RREF Voltage RREF Voltage Line Regulation RREF Pin Bias Current IREF Reference Current RILIM = 10k VC = 0V ISW = 0.5A VIN = 3V
The l denotes the specifications which apply over the full operating temperature range, otherwise specifications are at TA = 25C. VIN = 12V, unless otherwise noted.
CONDITIONS
l
MIN 3
TYP 3.5 0 7
MAX 40 1 1.29 3
UNITS V mA A A V A V kHz
SS = 0V VSHDN/UVLO = 0V SS = 0.4V UVLO Pin Voltage Rising VUVLO = 1V
l
1.15 2
1.22 2.5 0.7 1000
1.25 140 1.21 1.20
1.55 200 150 1.23 0.01
1.85 260 250 1.25 1.25 0.03 600
A mA mV V %/ V nA A
l l
3V < VIN < 40V (Note 3) Measured at RFB Pin with RREF = 6.49k
100 190
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LT3573 ELECTRICAL CHARACTERISTICS
PARAMETER Error Amplifier Voltage Gain Error Amplifier Transconductance Minimum Switching Frequency TC Current into RREF BIAS Pin Voltage VIN = 3V I = 10A, VIN = 3V VC = 0.35V RTC = 20.1k IBIAS = 30mA 2.9
The l denotes the specifications which apply over the full operating temperature range, otherwise specifications are at TA = 25C. VIN = 12V, unless otherwise noted.
CONDITIONS MIN TYP 150 150 40 27.5 3 3.1 MAX UNITS V/V mhos kHz A V
Note 1: Stresses beyond those listed under Absolute Maximum Ratings may cause permanent damage to the device. Exposure to any Absolute Maximum Rating condition for extended periods may affect device reliability and lifetime. Note 2: The LT3573E is guaranteed to meet performance specifications from 0C to 125C junction temperature. Specifications over the -40C
to 125C operating junction temperature range are assured by design characterization and correlation with statistical process controls. The LT3573I is guaranteed over the full -40C to 125C operating junction temperature range. Note 3: Current flows out of the RREF pin.
TYPICAL PERFORMANCE CHARACTERISTICS
Output Voltage
5.20 5.15 5.10 VOUT (V) 5.05 IQ (mA) 4 3 2 1 0 -50 VIN = 5V 5.00 4.95 4.90 4.85 4.80 -50 -25 50 25 0 75 TEMPERATURE (C) 100 125 7 6 5 VIN = 40V
TA = 25C, unless otherwise noted.
Quiescent Current
3.2 3.0 BIAS VOLTAGE (V) 2.8 2.6 2.4 2.2
Bias Pin Voltage
VIN = 40V VIN = 12V
-25
50 25 0 75 TEMPERATURE (C)
100
125
2.0 -50
-25
50 25 0 75 TEMPERATURE (C)
100
125
3573 G01
3573 G02
3573 G03
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LT3573 TYPICAL PERFORMANCE CHARACTERISTICS
Switch VCESAT
400 350 SWITCH VCESAT VOLTAGE (mV) CURRENT LIMIT (A) 300 250 200 150 100 50 0 0 0.25 0.50 0.75 1.00 SWITCH CURRENT (A) 1.25 1.50 125C -50C 25C 1.8 MAXIMUM CURRENT LIMIT 1.6 1.4 1.2 1.0 RILIM = 10k 0.8 0.6 0.4 MINIMUM CURRENT LIMIT 0.2 0 -50 -25 0 25 50 75 100 125 0.2 0 0 10 20 30 40 50
3573 G06
TA = 25C, unless otherwise noted.
Switch Current Limit
1.8 1.6 1.4 CURRENT LIMIT (A) 1.2 1.0 0.8 0.6 0.4
Switch Current Limit vs RILIM
TEMPERATURE (C)
3573 G04 3573 G05
RILIM RESISTANCE (k)
SHDN/UVLO Falling Threshold
1.28 12 10 SS PIN CURRENT (A)
3573 G07
SS Pin Current
SHDN/UVLO VOLTAGE (V)
1.26
8 6 4 2 0 -60 -40 -20 0 20 40 60 80 100 120 140 TEMPERATURE (C)
3573 G08
1.24
1.22
1.20
1.18 -60 -40 -20 0 20 40 60 80 100 120 140 TEMPERATURE (C)
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LT3573 PIN FUNCTIONS
GND: Ground. TEST: This pin is used for testing purposes only and must be connected to ground for the part to operate properly. SW: Collector Node of the Output Switch. This pin has large currents flowing through it. Keep the traces to the switching components as short as possible to minimize electromagnetic radiation and voltage spikes. VIN : Input Voltage. This pin supplies current to the internal start-up circuitry and as a reference voltage for the DCM comparator and feedback circuitry. This pin must be locally bypassed with a capacitor. BIAS: Bias Voltage. This pin supplies current to the switch driver and internal circuitry of the LT3573. This pin must be locally bypassed with a capacitor. This pin may also be connected to VIN if a third winding is not used and if VIN 15V. If a third winding is used, the BIAS voltage should be lower than the input voltage for proper operation. SHDN/UVLO: Shutdown/Undervoltage Lockout. A resistor divider connected to VIN is tied to this pin to program the minimum input voltage at which the LT3573 will operate. At a voltage below ~0.7V, the part draws no quiescent current. When below 1.25V and above ~0.7V, the part will draw 10A of current, but internal circuitry will remain off. Above 1.25V, the internal circuitry will start and a 10A current will be fed into the SS pin. When this pin falls below 1.25V, 2.5A will be pulled from the pin to provide programmable hysteresis for UVLO. RILIM: Maximum Current Limit Adjust Pin. A resistor should be tied to this pin to ground to set the current limit. Use a 10k resistor for the full current capabilities of the switch. SS: Soft-Start Pin. Place a soft-start capacitor here to limit start-up inrush current and output voltage ramp rate. Switching starts when the voltage at this pin reaches ~0.7V. VC: Compensation Pin for Internal Error Amplifier. Connect a series RC from this pin to ground to compensate the switching regulator. A 100pF capacitor in parallel helps eliminate noise. RFB: Input Pin for External Feedback Resistor. This pin is connected to the transformer primary (VSW). The ratio of this resistor to the RREF resistor, times the internal bandgap reference, determines the output voltage (plus the effect of any non-unity transformer turns ratio). The average current through this resistor during the flyback period should be approximately 200A. For nonisolated applications, this pin should be connected to VIN. RREF : Input Pin for External Ground-Referred Reference Resistor. This resistor should be in the range of 6k, but for convenience, need not be precisely this value. For nonisolated applications, a traditional resistor voltage divider may be connected to this pin. TC: Output Voltage Temperature Compensation. Connect a resistor to ground to produce a current proportional to absolute temperature to be sourced into the RREF node. ITC = 0.55V/RTC . Exposed Pad: Ground. The Exposed Pad of the package provides both electrical contact to ground and good thermal contact to the printed circuit board. The Exposed Pad must be soldered to the circuit board for proper operation and should be well connected with many vias to an internal ground plane.
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LT3573 BLOCK DIAGRAM
D1 VIN C1 R3 N:1 VIN TC CURRENT Q3 TC Q2 RFB FLYBACK ERROR AMP ONE SHOT SW L1A T1 L1B VOUT + C2 VOUT -
R6 RREF R4 BIAS C5 R1 1.22V SHDN/UVLO
I2 20A
1.22V
- gm +
CURRENT COMPARATOR
A2
- +
+
VIN
- A1 +
S S MASTER LATCH R Q
V1 120mV DRIVER BIAS
-
Q1
A4
+ -
RSENSE 0.02 GND
+ A5 -
R2
2.5A Q4 SS
INTERNAL REFERENCE AND REGULATORS
I1 7A
OSCILLATOR
VC R7 RILIM C3 R5
3573 BD
C4
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LT3573 OPERATION
The LT3573 is a current mode switching regulator IC designed specifically for the isolated flyback topology. The special problem normally encountered in such circuits is that information relating to the output voltage on the isolated secondary side of the transformer must be communicated to the primary side in order to maintain regulation. Historically, this has been done with optoisolators or extra transformer windings. Optoisolator circuits waste output power and the extra components increase the cost and physical size of the power supply. Optoisolators can also exhibit trouble due to limited dynamic response, nonlinearity, unit-to-unit variation and aging over life. Circuits employing extra transformer windings also exhibit deficiencies. Using an extra winding adds to the transformer's physical size and cost, and dynamic response is often mediocre. The LT3573 derives its information about the isolated output voltage by examining the primary side flyback pulse waveform. In this manner, no optoisolator nor extra transformer winding is required for regulation. The output voltage is easily programmed with two resistors. Since this IC operates in boundary control mode, the output voltage is calculated from the switch pin when the secondary current is almost zero. This method improves load regulation without external resistors and capacitors. The Block Diagram shows an overall view of the system. Many of the blocks are similar to those found in traditional switching regulators including: internal bias regulator, oscillator, logic, current amplifier and comparator, driver, and output switch. The novel sections include a special flyback error amplifier and a temperature compensation circuit. In addition, the logic system contains additional logic for boundary mode operation, and the sampling error amplifier. The LT3573 features a boundary mode control method, where the part operates at the boundary between continuous conduction mode and discontinuous conduction mode. The VC pin controls the current level just as it does in normal current mode operation, but instead of turning the switch on at the start of the oscillator period, the part detects when the secondary side winding current is zero. Boundary Mode Operation Boundary mode is a variable frequency, current-mode switching scheme. The switch turns on and the inductor current increases until a VC pin controlled current limit. The voltage on the SW pin rises to the output voltage divided by the secondary-to-primary transformer turns ratio plus the input voltage. When the secondary current through the diode falls to zero, the SW pin voltage falls below VIN . A discontinuous conduction mode (DCM) comparator detects this event and turns the switch back on. Boundary mode returns the secondary current to zero every cycle, so the parasitic resistive voltage drops do not cause load regulation errors. Boundary mode also allows the use of a smaller transformer compared to continuous conduction mode and no subharmonic oscillation. At low output currents the LT3573 delays turning on the switch, and thus operates in discontinuous mode. Unlike a traditional flyback converter, the switch has to turn on to update the output voltage information. Below 0.6V on the VC pin, the current comparator level decreases to its minimum value, and the internal oscillator frequency decreases in frequency. With the decrease of the internal oscillator, the part starts to operate in DCM. The output current is able to decrease while still allowing a minimum switch off-time for the error amp sampling circuitry. The typical minimum internal oscillator frequency with VC equal to 0V is 40kHz.
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LT3573 APPLICATIONS INFORMATION
ERROR AMPLIFIER--PSEUDO DC THEORY In the Block Diagram, the RREF (R4) and RFB (R3) resistors can be found. They are external resistors used to program the output voltage. The LT3573 operates much the same way as traditional current mode switchers, the major difference being a different type of error amplifier which derives its feedback information from the flyback pulse. Operation is as follows: when the output switch, Q1, turns off, its collector voltage rises above the VIN rail. The amplitude of this flyback pulse, i.e., the difference between it and VIN, is given as: VFLBK = (VOUT + VF + ISEC * ESR) * NPS VF = D1 forward voltage ISEC = Transformer secondary current ESR = Total impedance of secondary circuit NPS = Transformer effective primary-to-secondary turns ratio The flyback voltage is then converted to a current by the action of RFB and Q2. Nearly all of this current flows through resistor RREF to form a ground-referred voltage. This voltage is fed into the flyback error amplifier. The flyback error amplifier samples this output voltage information when the secondary side winding current is zero. The error amplifier uses a bandgap voltage, 1.23V, as the reference voltage. The relatively high gain in the overall loop will then cause the voltage at the RREF resistor to be nearly equal to the bandgap reference voltage VBG. The relationship between VFLBK and VBG may then be expressed as: V V FLBK = BG RFB RREF or, programming resistors, transformer turns ratio and diode forward voltage drop: R 1 VOUT = VBG FB - VF - ISEC (ES R) RREF NPS Additionally, it includes the effect of nonzero secondary output impedance (ESR). This term can be assumed to be zero in boundary control mode. More details will be discussed in the next section. Temperature Compensation The first term in the VOUT equation does not have a temperature dependence, but the diode forward drop has a significant negative temperature coefficient. To compensate for this, a positive temperature coefficient current source is connected to the RREF pin. The current is set by a resistor to ground connected to the TC pin. To cancel the temperature coefficient, the following equation is used: 1 VF R * = - FB * T R TC NPS 1 -RFB * R TC = NPS VF / T VTC or, T V R * TC FB T NPS
(VF /T) = Diode's forward voltage temperature coefficient (VTC /T) = 2mV VTC = 0.55V The resistor value given by this equation should also be verified experimentally, and adjusted if necessary to achieve optimal regulation over temperature. The revised output voltage is as follows: R 1 VOUT = VBG FB - VF RREF NPS V R - TC * FB - ISEC (ESR) R TC NPS
R 1 VFLBK = VBG FB RREF = Ratio of Q1 IC to IE, typically 0.986 VBG = Internal bandgap reference In combination with the previous VFLBK expression yields an expression for VOUT, in terms of the internal reference,
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LT3573 APPLICATIONS INFORMATION
ERROR AMPLIFIER--DYNAMIC THEORY Due to the sampling nature of the feedback loop, there are several timing signals and other constraints that are required for proper LT3573 operation. Minimum Current Limit The LT3573 obtains output voltage information from the SW pin when the secondary winding conducts current. The sampling circuitry needs a minimum amount of time to sample the output voltage. To guarantee enough time, a minimum inductance value must be maintained. The primary side magnetizing inductance must be chosen above the following value: 1 . 4H t L PRI VOUT * MIN * NPS = VOUT * NPS * IMIN V tMIN = minimum off-time, 350ns IMIN = minimum current limit, 250mA The minimum current limit is higher than that on the Electrical Characteristics table due to the overshoot caused by the comparator delay. Leakage Inductance Blanking When the output switch first turns off, the flyback pulse appears. However, it takes a finite time until the transformer primary side voltage waveform approximately represents the output voltage. This is partly due to the rise time on the SW node, but more importantly due to the transformer leakage inductance. The latter causes a very fast voltage spike on the primary side of the transformer that is not directly related to output voltage (some time is also required for internal settling of the feedback amplifier circuitry). The leakage inductance spike is largest when the power switch current is highest. In order to maintain immunity to these phenomena, a fixed delay is introduced between the switch turn-off command and the beginning of the sampling. The blanking is internally set to 150ns. In certain cases, the leakage inductance may not be settled by the end of the blanking period, but will not significantly affect output regulation. Selecting RFB and RREF Resistor Values The expression for VOUT, developed in the Operation section, can be rearranged to yield the following expression for RFB: RFB = where, VOUT = Output voltage VF = Switching diode forward voltage = Ratio of Q1, IC to IE, typically 0.986 NPS = Effective primary-to-secondary turns ratio VTC = 0.55V The equation assumes the temperature coefficients of the diode and VTC are equal, which is a good first-order approximation. Strictly speaking, the above equation defines RFB not as an absolute value, but as a ratio of RREF. So the next question is, "What is the proper value for RREF?" The answer is that RREF should be approximately 6.04k. The LT3573 is trimmed and specified using this value of RREF. If the impedance of RREF varies considerably from 6.04k, additional errors will result. However, a variation in RREF of several percent is acceptable. This yields a bit of freedom in selecting standard 1% resistor values to yield nominal RFB /RREF ratios. Tables 1-4 are useful for selecting the resistor values for RREF and RFB with no equations. The tables provide RFB, RREF and RTC values for common output voltages and common winding ratios.
Table 1. Common Resistor Values for 1:1 Transformers
VOUT (V) 3.3 5 12 15 20 NPS 1.00 1.00 1.00 1.00 1.00 RFB (k) 18.7 27.4 64.9 80.6 107 RREF (k) 6.04 6.04 6.04 6.04 6.04 RTC (k) 19.1 28 66.5 80.6 105
RREF * NPS ( VOUT + VF ) + VTC VBG
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LT3573 APPLICATIONS INFORMATION
Table 2. Common Resistor Values for 2:1 Transformers
VOUT (V) 3.3 5 12 15 NPS 2.00 2.00 2.00 2.00 RFB (k) 37.4 56 130 162 RREF (k) 6.04 6.04 6.04 6.04 RTC (k) 18.7 28 66.5 80.6
Table 3. Common Resistor Values for 3:1 Transformers
VOUT (V) 3.3 5 10 NPS 3.00 3.00 3.00 RFB (k) 56.2 80.6 165 RREF (k) 6.04 6.04 6.04 RTC (k) 20 28.7 54.9
relatively constant maximum output current regardless of input voltage. This is due to the continuous nonswitching behavior of the two currents. A flyback converter has both discontinuous input and output currents which makes it similar to a nonisolated buck-boost. The duty cycle will affect the input and output currents, making it hard to predict output power. In addition, the winding ratio can be changed to multiply the output current at the expense of a higher switch voltage. The graphs in Figures 1-3 show the maximum output power possible for the output voltages 3.3V, 5V, and 12V. The maximum power output curve is the calculated output power if the switch voltage is 50V during the off-time. To achieve this power level at a given input, a winding ratio value must be calculated to stress the switch to 50V, resulting in some odd ratio values. The curves below are examples of common winding ratio values and the amount of output power at given input voltages. One design example would be a 5V output converter with a minimum input voltage of 20V and a maximum input voltage of 30V. A three-to-one winding ratio fits this design example perfectly and outputs close to six watts at 30V but lowers to five watts at 20V.
Table 4. Common Resistor Values for 4:1 Transformers
VOUT (V) 3.3 5 NPS 4.00 4.00 RFB (k) 76.8 113 RREF (k) 6.04 6.04 RTC (k) 19.1 28
Output Power A flyback converter has a complicated relationship between the input and output current compared to a buck or a boost. A boost has a relatively constant maximum input current regardless of input voltage and a buck has a
8 7 OUTPUT POWER (W) 6 N = 3:1 5 4 3 2 1 0 0 5 10 15 20 25 30 35 40 45
3573 F01
MAXIMUM OUTPUT POWER N = 2:1 OUTPUT POWER (W)
8 7 6 5 N = 7:1 4 N = 2:1 3 2 1 0 0 5 10 15 20 25 N = 3:1 N = 5:1
8 MAXIMUM OUTPUT POWER OUTPUT POWER (W) 7 6 5 4 3 2 1 0 30 35 40 45 0 5 10 15 20 25 30 35 40 45
3573 F03
N = 2:1
MAXIMUM OUTPUT POWER
N = 3:1 N = 1:1
N = 1:1
INPUT VOLTAGE (V)
INPUT VOLTAGE (V)
3573 F02
INPUT VOLTAGE (V)
Figure 1. Output Power for 3.3V Output
Figure 2. Output Power for 5V Output
Figure 3. Output Power for 12V Output
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LT3573 APPLICATIONS INFORMATION
TRANSFORMER DESIGN CONSIDERATIONS Transformer specification and design is perhaps the most critical part of successfully applying the LT3573. In addition to the usual list of caveats dealing with high frequency isolated power supply transformer design, the following information should be carefully considered. Linear Technology has worked with several leading magnetic component manufacturers to produce pre-designed flyback transformers for use with the LT3573. Table 5 shows the details of several of these transformers.
Table 5. Predesigned Transformers--Typical Specifications, Unless Otherwise Noted
TRANSFORMER PART NUMBER PA2364NL PA2363NL PA2362NL PA2454NL PA2455NL PA2456NL SIZE (W x L x H) (mm) 15.24 x 13.1 x 11.45 15.24 x 13.1 x 11.45 15.24 x 13.1 x 11.45 15.24 x 13.1 x 11.45 15.24 x 13.1 x 11.45 15.24 x 13.1 x 11.45 LPRI (H) 25 25 24 24 25 25 LLEAKAGE (nH) 1000 850 550 430 450 390 NP:NS:NB 7:1:1 5:1:1 4:1:1 3:1:1 2:1:1 1:1:1 RPRI (m) 125 117 117 82 82 82 RSEC (m) 5.6 7.5 9.5 11 22 84 VENDOR Pulse Engineering Pulse Engineering Pulse Engineering Pulse Engineering Pulse Engineering Pulse Engineering TARGET APPLICATIONS 12V- >3.3V, 1.5A 12V- >5V, 1A 24V- >3.3V, 1.5A 24V- >5V, 1A 24V- >12V, 0.5A 12V- >12V, 0.3A 24V- >12V, 0.4A 36V- >5V, 0.6A 24V- >15V, 0.4A 24V- >5V, 1A 24V- >5V, 1A 24V- >3.3V, 1.5A 12V- >5V, 1A 12V- >3.3V, 1.5A 24V- >5V, 1A 24V- >3.3V, 1.5A 24V- >12V, 0.5A 24V- >12V, 0.3A 24V- >12V, 0.4A 36V- >5V, 0.6A 24V- >5V, 0.5A 24V- >15V, 0.4A 24V- >5V, 1A 24V- >5V, 1A 24V- >5V, 1A 24V- >5V, 1A 5V- >5V, 0.2A
PA2617NL PA2626NL PA2627NL GA3429-BL GA3430-BL GA3431-BL 750310471 750310559 750310562 750310563
12.70 x 10.67 x 9.14 12.70 x 10.67 x 9.14 15.24 x 13.1 x 11.45 15.24 x 12.7 x 11.43 15.24 x 12.7 x 11.43 15.24 x 12.7 x 11.43 15.24 x 13.3 x 11.43 15.24 x 13.3 x 11.43 15.24 x 13.3 x 11.43 15.24 x 13.3 x 11.43
21 30 50 25 25 25 25 24 25 25
245 403 766 566 685 945 350 400 330 325
1:1:0.33 3:1:1 3:1:1 4:1:1 5:1:1 7:1:1 3:1:1 4:1:1 2:1:1 1:1:0.5
164 240 420 95 90 90 57 51 60 60
166 66 44 7.5 5.5 5.5 11 16 20 60
Pulse Engineering Pulse Engineering Pulse Engineering Coilcraft Coilcraft Coilcraft Wurth Elektronik Wurth Elektronik Wurth Elektronik Wurth Elektronik
750310564 750310799 750370040 750370041 750370047 L11-0059 L10-1019
15.24 x 13.3 x 11.43 9.14 x 9.78 x 10.54 9.14 x 9.78 x 10.54 9.14 x 9.78 x 10.54 13.35 x 10.8 x 9.14 9.52 x 9.52 x 4.95 9.52 x 9.52 x 4.95
63 25 30 50 30 24 18
450 125 150 450 150
3:1:1 1:1:0.33 3:1:1 3:1:1 3:1:1 3:1 1:1
115 60 60 190 60 266 90
50 74 12.5 26 12.5 266 90
Wurth Elektronik Wurth Elektronik Wurth Elektronik Wurth Elektronik Wurth Elektronik BH Electronics BH Electronics
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LT3573 APPLICATIONS INFORMATION
Turns Ratio Note that when using an RFB /RREF resistor ratio to set output voltage, the user has relative freedom in selecting a transformer turns ratio to suit a given application.In contrast, simpler ratios of small integers, e.g., 1:1, 2:1, 3:2, etc., can be employed to provide more freedom in setting total turns and mutual inductance. Typically, the transformer turns ratio is chosen to maximize available output power. For low output voltages (3.3V or 5V), a N:1 turns ratio can be used with multiple primary windings relative to the secondary to maximize the transformer's current gain (and output power). However, remember that the SW pin sees a voltage that is equal to the maximum input supply voltage plus the output voltage multiplied by the turns ratio. This quantity needs to remain below the ABS MAX rating of the SW pin to prevent breakdown of the internal power switch. Together these conditions place an upper limit on the turns ratio, N, for a given application. Choose a turns ratio low enough to ensure: 50 V - VIN(MAX ) N< VOUT + VF For larger N:1 values, a transformer with a larger physical size is needed to deliver additional current and provide a large enough inductance value to ensure that the off-time is long enough to accurately measure the output voltage. For lower output power levels, a 1:1 or 1:N transformer can be chosen for the absolute smallest transformer size. A 1:N transformer will minimize the magnetizing inductance (and minimize size), but will also limit the available output power. A higher 1:N turns ratio makes it possible to have very high output voltages without exceeding the breakdown voltage of the internal power switch. Linear Technology has worked with several magnetic component manufacturers to produce predesigned flyback transformers for use with the LT3573. Table 5 shows the details of several of these transformers. Leakage Inductance Transformer leakage inductance (on either the primary or secondary) causes a voltage spike to appear at the primary after the output switch turns off. This spike is increasingly prominent at higher load currents where more stored energy must be dissipated. In most cases, a snubber circuit will be required to avoid overvoltage breakdown at the output switch node. Transformer leakage inductance should be minimized. An RCD (resistor capacitor diode) clamp, shown in Figure 4, is required for most designs to prevent the leakage inductance spike from exceeding the breakdown voltage of the power device. The flyback waveform is depicted in Figure 5. In most applications, there will be a very fast voltage spike caused by a slow clamp diode that may not exceed 60V. Once the diode clamps, the leakage inductance current is absorbed by the clamp capacitor. This period should not last longer than 150ns so as not to interfere with the output regulation, and the voltage during this clamp period must not exceed 55V. The clamp diode turns off after the leakage inductance energy is absorbed and the switch voltage is then equal to: VSW(MAX) = VIN(MAX) + N(VOUT + VF) This voltage must not exceed 50V. This same equation also determines the maximum turns ratio. When choosing the snubber network diode, careful attention must be paid to maximum voltage seen by the SW pin. Schottky diodes are typically the best choice to be used in the snubber, but some PN diodes can be used if they turn on fast enough to limit the leakage inductance spike. The leakage spike must always be kept below 60V. Figures 6 and 7 show the SW pin waveform for a 24VIN, 5VOUT application at a 1A load current. Notice that the leakage spike is very high (more than 65V) with the "bad" diode, while the "good" diode effectively limits the spike to less than 55V.
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LT3573 APPLICATIONS INFORMATION
LS - + C R
VSW < 60V < 55V < 50V
D
t OFF > 350ns tSP < 150ns
3573 F05
TIME
3573 F04
Figure 4. RCD Clamp
Figure 5. Maximum Voltages for SW Pin Flyback Waveform
10V/DIV
10V/DIV
100ns/DIV
3573 F06
100ns/DIV
3573 F07
Figure 6. Good Snubber Diode Limits SW Pin Voltage
Figure 7. Bad Snubber Diode Does Not Limit SW Pin Voltage
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13
LT3573 APPLICATIONS INFORMATION
Secondary Leakage Inductance In addition to the previously described effects of leakage inductance in general, leakage inductance on the secondary in particular exhibits an additional phenomenon. It forms an inductive divider on the transformer secondary that effectively reduces the size of the primary-referred flyback pulse used for feedback. This will increase the output voltage target by a similar percentage. Note that unlike leakage spike behavior, this phenomenon is load independent. To the extent that the secondary leakage inductance is a constant percentage of mutual inductance (over manufacturing variations), this can be accommodated by adjusting the RFB /RREF resistor ratio. Winding Resistance Effects Resistance in either the primary or secondary will reduce overall efficiency (POUT /PIN). Good output voltage regulation will be maintained independent of winding resistance due to the boundary mode operation of the LT3573. Bifilar Winding A bifilar, or similar winding technique, is a good way to minimize troublesome leakage inductances. However, remember that this will also increase primary-to-secondary capacitance and limit the primary-to-secondary breakdown voltage, so bifilar winding is not always practical. The Linear Technology applications group is available and extremely qualified to assist in the selection and/or design of the transformer. Setting the Current Limit Resistor The maximum current limit can be set by placing a resistor between the RILIM pin and ground. This provides some flexibility in picking standard off-the-shelf transformers that may be rated for less current than the LT3573's internal power switch current limit. If the maximum current limit is needed, use a 10k resistor. For lower current limits, the following equation sets the approximate current limit: RILIM = 65 * 10 3(1 . 6 A - ILIM ) + 10k
LT3573 R2
The Switch Current Limit vs RILIM plot in the Typical Performance Characteristics section depicts a more accurate current limit. Undervoltage Lockout (UVLO) The SHDN/UVLO pin is connected to a resistive voltage divider connected to VIN as shown in Figure 8. The voltage threshold on the SHDN/UVLO pin for VIN rising is 1.22V. To introduce hysteresis, the LT3573 draws 2.5A from the SHDN/UVLO pin when the pin is below 1.22V. The hysteresis is therefore user-adjustable and depends on the value of R1. The UVLO threshold for VIN rising is: VIN(UVLO,RISING) = 1 . 22V * (R1 + R2) + 2 . 5A * R1 R2
The UVLO threshold for VIN falling is: VIN(UVLO,FALLING) = 1 . 22V * (R1 + R2) R2
To implement external run/stop control, connect a small NMOS to the UVLO pin, as shown in Figure 8. Turning the NMOS on grounds the UVLO pin and prevents the LT3573 from operating, and the part will draw less than a 1A of quiescent current.
VIN R1 SHDN/UVLO RUN/STOP CONTROL (OPTIONAL)
GND
3573 F08
Figure 8. Undervoltage Lockout (UVLO)
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14
LT3573 APPLICATIONS INFORMATION
Minimum Load Requirement The LT3573 obtains output voltage information through the transformer while the secondary winding is conducting current. During this time, the output voltage (multiplied times the turns ratio) is presented to the primary side of the transformer. The LT3573 uses this reflected signal to regulate the output voltage. This means that the LT3573 must turn on every so often to sample the output voltage, which delivers a small amount of energy to the output. This sampling places a minimum load requirement on the output of 1% to 2% of the maximum load. BIAS Pin Considerations For applications with an input voltage less than 15V, the BIAS pin is typically connected directly to the VIN pin. For input voltages greater than 15V, it is preferred to leave the BIAS pin separate form the VIN pin. In this condition, the BIAS pin is regulated with an internal LDO to a voltage of 3V. By keeping the BIAS pin separate from the input voltage at high input voltages, the physical size of the capacitors can be minimized (the BIAS pin can then use a 6.3V or 10V rated capacitor). Overdriving the BIAS Pin with a Third Winding The LT3573 provides excellent output voltage regulation without the need for an optocoupler, or third winding, but for some applications with higher input voltages (>20V), it may be desirable to add an additional winding (often called a third winding) to improve the system efficiency. For proper operation of the LT3573, if a winding is used as a supply for the BIAS pin, ensure that the BIAS pin voltage is at least 3.15V and always less than the input voltage. For a typical 24VIN application, overdriving the BIAS pin will improve the efficiency gain 4-5%. Loop Compensation The LT3573 is compensated using an external resistorcapacitor network on the VC pin. Typical values are in the range of RC = 50k and CC = 1nF (see the numerous schematics in the Typical Applications section for other possible values). If too large of an RC value is used, the part will be more susceptible to high frequency noise and jitter. If too small of an RC value is used, the transient performance will suffer. The value choice for CC is somewhat the inverse of the RC choice: if too small a CC value is used, the loop may be unstable, and if too large a CC value is used, the transient performance will also suffer. Transient response plays an important role for any DC/DC converter. Design Example The following example illustrates the converter design process using LT3573. Given the input voltage of 20V to 28V, the required output is 5V, 1A. VIN(MIN) = 20V, VIN(MAX) = 28V, VOUT = 5V, VF = 0.5V and IOUT = 1A 1. Select the transformer turns ratio to accommodate the output. The output voltage is reflected to the primary side by a factor of turns ratio N. The switch voltage stress VSW is expressed as: N N= P NS VSW(MAX ) = VIN + N( VOUT + VF ) < 50 V Or rearranged to: 50 - VIN(MAX ) N< ( VOUT + VF ) On the other hand, the primary side current is multiplied by the same factor of N. The converter output capability is: 1 IOUT(MAX ) = 0 . 8 * (1 - D) * NI PK 2 N( VOUT + VF ) D= VIN + N( VOUT + VF )
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15
LT3573 APPLICATIONS INFORMATION
The transformer turns ratio is selected such that the converter has adequate current capability and a switch stress below 50V. Table 6 shows the switch voltage stress and output current capability at different transformer turns ratio.
Table 6. Switch Voltage Stress and Output Current Capability vs Turns-Ratio
N 1:1 2:1 3:1 4:1 VSW(MAX) AT VIN(MAX) (V) 33.5 39 44.5 50 IOUT(MAX) AT VIN(MIN) (A) 0.53 0.88 1.12 1.30 DUTY CYCLE (%) 16~22 28~35 37~45 44~52
Table 7.Switching Frequency at Different Primary Inductance at IPK
L (H) 25 50 100 fSW AT VIN(MIN) (kHz) 236 121 61 fSW AT VIN(MAX) (kHz) 305 157 80
Note: The switching frequency is calculated at maximum output.
In this design example, the minimum primary inductance is used to achieve a nominal switching frequency of 275kHz at full load. The PA2454NL from Pulse Engineering is chosen as the flyback transformer. Given the turns ratio and primary inductance, a customized transformer can be designed by magnetic component manufacturer or a multi-winding transformer such as a Coiltronics Versa-Pac may be used. 3. Select the output diodes and output capacitor. The output diode voltage stress VD is the summation of the output voltage and reflection of input voltage to the secondary side. The average diode current is the load current. VD = VOUT + VIN N
BIAS winding turns ratio is selected to program the BIAS voltage to 3~5V. The BIAS voltage shall not exceed the input voltage. The turns ratio is then selected as primary: secondary: BIAS = 3:1:1. 2. Select the transformer primary inductance for target switching frequency. The LT3573 requires a minimum amount of time to sample the output voltage during the off-time. This off-time, tOFF(MIN), shall be greater than 350ns over all operating conditions. The converter also has a minimum current limit, LMIN, of 250mA to help create this off-time. This defines the minimum required inductance as defined as: N( VOUT + VF ) L MIN = * t OFF(MIN) IMIN The transformer primary inductance also affects the switching frequency which is related to the output ripple. If above the minimum inductance, the transformer's primary inductance may be selected for a target switching frequency range in order to minimize the output ripple. The following equation estimates the switching frequency. 1 1 fSW = = IPK IPK t ON + t OFF + VIN NPS ( VOUT + VF ) L L
The output capacitor should be chosen to minimize the output voltage ripple while considering the increase in size and cost of a larger capacitor. The following equation calculates the output voltage ripple. LI 2PK VMAX = 2 CVOUT 4. Select the snubber circuit to clamp the switch voltage spike. A flyback converter generates a voltage spike during switch turn-off due to the leakage inductance of the transformer. In order to clamp the voltage spike below the maximum rating of the switch, a snubber circuit is used. There are many types of snubber circuits, for example R-C, R-C-D
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16
LT3573 APPLICATIONS INFORMATION
and Zener clamps. Among them, RCD is widely used. Figure 9 shows the RCD snubber in a flyback converter. A typical switch node waveform is shown in Figure 10. During switch turn-off, the energy stored in the leakage inductance is transferred to the snubber capacitor, and eventually dissipated in the snubber resistor. V ( V - N * VOUT ) 1 L S I2PK fSW = C C 2 R The snubber resistor affects the spike amplitude VC and duration tSP, the snubber resistor is adjusted such that tSP is about 150ns. Prolonged tSP may cause distortion to the output voltage sensing. The previous steps finish the flyback power stage design. 5. Select the feedback resistor for proper output voltage. Using the resistor Tables 1-4, select the feedback resistor RFB, and program the output voltage to 5V. Adjust the RTC resistor for temperature compensation of the output voltage. RREF is selected as 6.04k. A small capacitor in parallel with RREF filters out the noise during the voltage spike, however, the capacitor should limit to 10pF A large capacitor causes distortion on volt. age sensing. 6. Optimize the compensation network to improve the transient performance. The transient performance is optimized by adjusting the compensation network. For best ripple performance, select a compensation capacitor not less than 1nF and select a , compensation resistor not greater than 50k. 7. Current limit resistor, soft-start capacitor and UVLO resistor divider Use the current limit resistor RLIM to lower the current limit if a compact transformer design is required. Soft-start capacitor helps during the start-up of the flyback converter. Select the UVLO resistor divider for intended input operation range. These equations are aforementioned.
LS - + C R
VC NVOUT
D
VIN
tSP
3573 F09
3573 F10
Figure 9. RCD Snubber in a Flyback Converter
Figure 10. Typical Switch Node Waveform
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17
LT3573 TYPICAL APPLICATIONS
Low Input Voltage 5V Isolated Flyback Converter
VIN 5V 3:1 C1 10F R1 200k SHDN/UVLO R2 90.9k LT3573 RFB RREF R4 6.04k TC RILIM SS VC R6 28.7k R5 10k C2 10nF SW SW GND TEST BIAS R7 57.6k C3 1000pF
3573 TA02
D1
VOUT+ 5V, 350mA C5 47F VOUT-
VIN
C6 0.22F
R8 T1 2k 24H D2
2.6H
R3 80.6k
VIN
T1: PULSE PA2454NL OR WURTH ELEKTRONIK 750310471 D1: B340A D2: 1N4148 C5: MURATA, GRM32ER71A476K
12V Isolated Flyback Converter
VIN 5V 2:1:1 C1 10F R1 200k SHDN/UVLO R2 90.9k LT3573 RFB RREF R4 6.04k TC RILIM SS VC R6 59k R5 10k C2 10nF SW SW GND TEST BIAS R7 56.2k C3 3300pF VIN T1: COILTRONICS VPH1-0076-R D1, D2: B240A D3: 1N4148 C5, C6: MURATA, GRM32ER71A476K R3 118k D3 D2 VOUT2+ 10.9H C6 47F VOUT2- -12V, 100mA C6 0.22F R8 T1 2k 43.6H D1 VOUT1+ 12V, 100mA C5 47F VOUT1-
VIN
10.9H
3573 TA03
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18
LT3573 TYPICAL APPLICATIONS
5V Isolated Flyback Converter
VIN 12V TO 24V (*40V) 3:1:1 C1 10F R1 499k SHDN/UVLO R2 71.5k LT3573 RFB RREF R4 6.04k TC RILIM SS VC R6 28.7k R5 10k R7 45.3k C3 1000pF GND TEST BIAS D2 C4 4.7F L1C 2.6H *OPTIONAL THIRD WINDING FOR 40V OPERATION SW R3 80.6k D3 C6 0.22F R8 T1 2k 24H D1 VOUT + 5V, 700mA C5 47F VOUT -
VIN
2.6H
C2 10nF
3573 TA04
T1: PULSE PA2454NL OR WURTH ELEKTRONIK 750370047 D1: B340A D3: 1N4148 C5: MURATA, GRM32ER71A476K
Efficiency
90 80 70 EFFICIENCY (%) 60 50 40 30 20 10 0 0 200 400 600 800 1000 1200 1400
3573 TA04b
VIN = 24V VIN = 12V
IOUT (mA)
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19
LT3573 TYPICAL APPLICATIONS
3.3V Isolated Flyback Converter
VIN 12V TO 24V (*40V) 4:1:1 C1 10F R1 499k SHDN/UVLO R2 71.5k LT3573 RFB RREF R4 6.04k TC RILIM SS VC R6 19.1k R5 10k C2 10nF
3573 TA05
D1
VOUT + 3.3V, 1A C5 47F VOUT -
VIN
C6 0.22F
R8 T1 2k 24H D3
1.5H
R3 76.8k
SW
GND TEST BIAS R7 25.5k C3 1500pF D2 C4 4.7F L1C 1.5H *OPTIONAL THIRD WINDING FOR 40V OPERATION
T1: PULSE PA2362NL OR COILCRAFT GA3429-BL D1: B340A D3: 1N4148
12V Isolated Flyback Converter
VIN 5V 3:1 C1 10F R1 499k SHDN/UVLO R2 71.5k LT3573 RFB RREF TC RILIM SS VC R6 59k R5 10k C2 10nF
3573 TA06
D1
VOUT 12V, 400mA C5 47F VOUT-
VIN
C6 0.22F
R8 T1 2k 58.5H D2
6.5H
R3 178k
R4 6.04k SW GND TEST BIAS R7 40.2k C3 4700pF VIN T1: COILTRONICS VP1-0102-R D1: B340A D2: 1N4148
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20
LT3573 TYPICAL APPLICATIONS
Four Output 12V Isolated Flyback Converter
VIN 12V TO 24V 2:1:1:1:1 C1 10F R1 499k SHDN/UVLO R2 71.5k LT3573 TC RILIM SS VC R6 59k R5 10k C2 10nF R7 20k C3 0.01F SW GND TEST BIAS VIN 10.9H RFB RREF R4 6.04k 10.9H D3 R3 118k VIN C6 0.22F T1 R8 2k 43.6H 10.9H D1 VOUT1+ 12V, 60mA C5 47F VOUT1- D5 10.9H D2 VOUT2+ 12V, 60mA VOUT2- VOUT3+ 12V, 60mA
C6 47F
C7 47F VOUT3-
D4
C8 47F
VOUT4+ 12V, 60mA VOUT4-
T1: COILTRONICS VPH1-0076-R D1-D4: B240A D5: 1N4148
3573 TA07
5V Isolated Flyback Converter Using a Tiny Transformer
D1
VIN 12V
3:1 C1 10F R1 200k SHDN/UVLO R2 90.9k LT3573 RFB RREF TC RILIM SS VC R6 28.7k R5 30k C2 10nF
3573 TA08
VOUT 5V, 600mA C5 47F VOUT-
VIN
C6 0.22F
R8 2k
T1 20H
2.2H
R3 80.6k
D2
R4 6.04k SW GND TEST BIAS R7 47.5k C3 1000pF VIN T1: BH ELECTRONICS L11-0059 D1: B340A D2: 1N4148
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21
LT3573 TYPICAL APPLICATIONS
5V Isolated Flyback Converter Using Coupling Inductor
VIN 5V C1 10F R1 200k SHDN/UVLO R2 90.9k LT3573 RFB RREF TC RILIM SS VC R6 26.1k R5 10k C2 10nF
3573 TA09
1:1 C6 0.22F R8 T1 2k 23.6H D2
D1
VOUT+ 5V, 0.2A C5 47F VOUT-
VIN
23.6H
R3 26.1k
R4 6.04k SW GND TEST BIAS R7 56.2k C3 1500pF VIN T1: BH ELECTRONICS, L10-1022 D1: B220A D2: CMD5H-3
300V Isolated Flyback Converter
VIN 5V TO 15V C1 10F R1 100k SHDN/UVLO R2 36k LT3573 RFB RREF TC RILIM SS VC R6 20.5k R5 10k C2 10nF
3573 TA10
1:17 C6 0.22F R8 4.7k T1 26H
D1
VOUT+ 300V, 5mA C5, 0.47F 400V , POLY FILM VOUT-
VIN
7.583mH
R3 90.9k
D2
R4 6.04k SW GND TEST BIAS R7 10k C3 1000pF VIN D1: TOSHIBA CRF02 D2: ZETEX ZHCS 506TA
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22
LT3573 PACKAGE DESCRIPTION
MSE Package 16-Lead Plastic MSOP, Exposed Die Pad
(Reference LTC DWG # 05-08-1667 Rev A)
BOTTOM VIEW OF EXPOSED PAD OPTION 2.845 (.112 0.102 .004) 0.889 (.035
0.127 .005)
2.845 (.112 1
0.102 .004) 8 0.35 REF
5.23 (.206) MIN
1.651 (.065
0.102 3.20 - 3.45 .004) (.126 - .136)
1.651 (.065
0.102 .004)
0.12 REF
0.305 0.038 (.0120 .0015) TYP
0.50 (.0197) BSC
16 4.039 0.102 (.159 .004) (NOTE 3)
DETAIL "B" CORNER TAIL IS PART OF DETAIL "B" THE LEADFRAME FEATURE. FOR REFERENCE ONLY 9 NO MEASUREMENT PURPOSE
RECOMMENDED SOLDER PAD LAYOUT
16151413121110 9
0.280 0.076 (.011 .003) REF
0.254 (.010) GAUGE PLANE
DETAIL "A" 0 - 6 TYP 4.90 0.152 (.193 .006) 3.00 0.102 (.118 .004) (NOTE 4)
0.53 0.152 (.021 .006) DETAIL "A" 0.18 (.007) 12345678 1.10 (.043) MAX 0.86 (.034) REF
SEATING PLANE
NOTE: 1. DIMENSIONS IN MILLIMETER/(INCH) 2. DRAWING NOT TO SCALE 3. DIMENSION DOES NOT INCLUDE MOLD FLASH, PROTRUSIONS OR GATE BURRS. MOLD FLASH, PROTRUSIONS OR GATE BURRS SHALL NOT EXCEED 0.152mm (.006") PER SIDE 4. DIMENSION DOES NOT INCLUDE INTERLEAD FLASH OR PROTRUSIONS. INTERLEAD FLASH OR PROTRUSIONS SHALL NOT EXCEED 0.152mm (.006") PER SIDE 5. LEAD COPLANARITY (BOTTOM OF LEADS AFTER FORMING) SHALL BE 0.102mm (.004") MAX
0.17 -0.27 (.007 - .011) TYP
0.50 (.0197) BSC
0.1016 (.004
0.0508 .002)
MSOP (MSE16) 0608 REV A
3573f
Information furnished by Linear Technology Corporation is believed to be accurate and reliable. However, no responsibility is assumed for its use. Linear Technology Corporation makes no representation that the interconnection of its circuits as described herein will not infringe on existing patent rights.
23
LT3573 TYPICAL APPLICATION
9V to 30VIN, +5V/-5VOUT Isolated Flyback Converter
VIN 9V TO 30V C1 10F R1 357k SHDN/UVLO R2 51.1k LT3573 RFB RREF TC RILIM SS VC R5 28.7k R6 10k C2 10nF SW GND TEST BIAS R7 23.7k C3 2700pF D3 C4 4.7F L1D 7H *OPTIONAL THIRD WINDING FOR >24V OPERATION T1: WURTH ELEKTRONIK 750310564
3573 TA11
T1 3:1:1:1 C6 0.22F R8 L1A 2k 63H D4 R3 80.6k L1B 7H
D1
VOUT + +5V, 350mA C5 47F COM
VIN
L1C 7H D2
C6 47F VOUT - -5V, 350mA
R4 6.04k
RELATED PARTS
PART NUMBER LT1424-5 LT1424-9 LT1425 LTC(R)1624 LT1725 LT1737 DESCRIPTION Isolated Flyback Switching Regulator Isolated Flyback Switching Regulator Isolated Flyback Switching Regulator with No External Power Devices Current Mode DC/DC Controller General Purpose Isolated Flyback Controller High Power Isolated Flyback Controller
TM
COMMENTS 5V Output Voltage, No Optoisolator Required 9V Output, Regulation Maintained Under Light Loads No Optoisolator or "Third Winding" Required, Up to 6W Output 300kHz Operating Frequency; Buck, Boost, SEPIC Topologies; VIN Up to 36V, SO-8 Package No Optoisolator Required, VIN and VOUT Limited Only by External Power Components No Optoisolator or "Third Winding" Required, Up to 50W Output Adjustable Switching Frequency, 2.5V VIN 36V, Optional Burst Mode(R) Operation at Light Load 550kHz Switching Frequency, 2.5V to 9.8V VIN Range Controller for Forward, Boost, Flyback and SEPIC Converters from 30W to 300W VIN and VOUT Limited Only by External Components VIN and VOUT Limited Only by External Components High Efficiency (89%); Multiple Output with Excellent Cross Regulation
LTC1871/LTC1871-1, Wide Input Range, No RSENSE Current Mode Flyback, Boost and SEPIC Controller LTC1871-7 LTC1872, LTC1872B SOT-23 Constant-Frequency Current Mode Boost DC/DC Controller LT1950 Current Mode PWM Controller LTC3803/LTC3803-5 200kHz Flyback DC/DC Controller LTC3805/LTC3805-5 Adjustable Frequency Flyback Controller LTC3806 LT3825 LT3837 LTC3872 Synchronous Flyback Controller
Isolated No-Opto Synchronous Flyback Controller VIN 16V to 75V Limited by External Components, Up to 80W, Current Mode Control Isolated No-Opto Synchronous Flyback Controller VIN 4.5V to 36V Limited by External Components, Up to 60W, Current Mode Control No RSENSE Boost Controller 550kHz Fixed Frequency, 2.75V VIN 9.8V, ThinSOTTM or DFN Package VIN and VOUT Limited by External Components, 200kHz Frequency, ThinSOT or DFN Package
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LTC3873/LTC3873-5 No RSENSE Constant-Frequency Boost/Flyback/ SEPIC Controller
Burst Mode is a registered trademark of Linear Technology Corporation. No RSENSE and ThinSOT are trademarks of Linear Technology Corporation.
24 Linear Technology Corporation
(408) 432-1900 FAX: (408) 434-0507
LT 1008 * PRINTED IN USA
1630 McCarthy Blvd., Milpitas, CA 95035-7417
www.linear.com
(c) LINEAR TECHNOLOGY CORPORATION 2008

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